The present invention relates to a matched filter for use in an inverse spread process of a received signal using a spreading signal such as a pseudo-noise (PN) signal in a spread spectrum receiver.
Conventionally, as the method of an inverse spread process in a spread spectrum receiver, two methods have been used. One is a passive correlation method which uses a matched filter as inverse spreading means and the other is an active correlation method which uses a correlator as inverse spreading means.
First, the operation of the matched filter will be described. FIG. 6 shows, as an example of the arrangement of the matched filter, a matched filter which uses a spreading signal as a coefficient in an N tap transversal type filter.
In FIG. 6, D(m) and R(m) indicate a received signal and a correlation signal at instant m, respectively. Also, in FIG. 6, N spreading signals P(n) (n=0, 1, . . . , Nxe2x88x922, Nxe2x88x921) indicate spreading signals of period N. The received signal D(m) is sampled with respect to time with the period equal to chip section length Tc, where Tc is a section length (chip section length) of the spreading signal P(n).
Note that, the value in the brackets of the spreading signal P(n) indicates time so that the larger the value, the further back from the present time. Namely, for example, comparing P(n) and P(n+1), P(n+1) indicates a signal which is further back from P(n) by the amount of time equal to the chip section length Tc. On the other hand, the value in the brackets of the other signals such as the received signal D(m) and correlation signal R(m) indicates time so that the smaller the value, the further back from the present time. Namely, for example, comparing D(m) and D(m+1), D(m) indicates a signal which is further back from D(m+1) by the amount of time equal to the chip section length Tc.
When the section length (symbol section length) of data to be subjected to a spread process on the sending side is Ts, the spread ratio N has the relationship of N=Ts/Tc with the chip section length Tc and symbol section length Ts.
As shown in FIG. 6, in a common matched filter, the tap number is equal to the spread ratio N. In the following, for convenience of explaining the operation, the received signal is regarded as a signal within a baseband.
A delay circuit 61 has the structure in which Nxe2x88x921 delay elements 61(p) (p=1, . . . , Nxe2x88x922, Nxe2x88x921) are connected in series. To the delay element 61(1) is inputted the received signal D(m). Each delay element 61(p) outputs signal D(m-p). The delay time of each delay element 61(p) is equal to the chip section length Tc.
In a multiplier 62(n), the spreading signal P(n) is multiplied with each of the input signal D(m) and the output signal D(m-p) of each delay element 61(p) (i.e., signal D(m-n)), and the result of this multiplication is outputted as a signal. Then, the output signals of all the multipliers 62(n) are added in an adder 63. As a result, a correlation signal R(m) with respect to symbol section length Ts, which is the length of one period of the spreading signal P(n), is determined. This is represented by the following equation (1).                               R          ⁡                      (            m            )                          =                              ∑                          n              =              0                                      N              -              1                                ⁢                                    D              ⁡                              (                                  m                  -                  n                                )                                      ·                          P              ⁡                              (                n                )                                                                        (        1        )            
A common spreading signal only takes the two values of +1 and xe2x88x921, and for this reason in a common multiplier 62(n), the polarity of the input of the adder 63 is reversed in accordance with the spreading signal P(n). As is clear from the structure of FIG. 6, in the matched filter, the spreading signal P(n) is fixed, and the cross-correlation function with the received signal D(m), which changes per chip section length Tc, is calculated. The absolute value of the correlation signal R(m) becomes maximum at the time when the received signal D(m) and the spreading signal P(n) are in-phase. By the periodicity of the received signal D(m) and the spreading signal P(n), the time of in-phase arrives per symbol section length Ts. Thus, it is ensured that the inverse spread using a matched filter is carried out with the period equal to the symbol section length Ts. Accordingly, the operation for making a coincidence of the phases of the received signal D(m) and the spreading signal P(n) is not required. For this reason, the inverse spread method using the matched filter is called the passive correlation method.
The following describes the operation of the correlator. FIG. 7 shows an example of the structure of the correlator. First, the products of the received signal D(m) and N spreading signals P(n) (n=0, 1, 2, . . . , Nxe2x88x921) are determined in the multiplier 64, and the products are integrated in an integrator 65. The received signal D(m) has been sampled with respect to time, and for this reason in this integration operation, cumulative addition of the products of the received signal D(m) and the spreading signals P(n) is carried out. The correlation signal R(m) of the correlator is represented by the following equation (2).                               R          ⁡                      (            m            )                          =                              ∑                          n              =              0                                      N              -              1                                ⁢                                    D              ⁡                              (                                  m                  -                  n                                )                                      ·                          P              (                                                (                                      n                    +                    i                                    )                                ⁢                                  xe2x80x83                                ⁢                mod                ⁢                                  xe2x80x83                                ⁢                N                            )                                                          (        2        )            
Here, i is an integer in the range of 0xe2x89xa6ixe2x89xa6Nxe2x88x921 indicating the phase of the spreading signal P(n) at the start of integration, and i is set by a controller 67. xe2x80x9cmodxe2x80x9d is residual operator.
A spreading signal generator 66 outputs the spreading signal P(n) per chip section length Tc. For example, when i=5, and Nxe2x89xa77, the spreading signal generator 66 outputs spreading signal P(4) at instant mxe2x88x92N+1, spreading signal P(3) at instant mxe2x88x92N+2, . . . , spreading signal P(0) at instant mxe2x88x92N+5, spreading signal P(Nxe2x88x921) at instant mxe2x88x92N+6, . . . , and spreading signal P(5) at instant m.
In this manner, in the above correlator, the integrator 65 performs integration of the product of the received signal D(m) and the spreading signal P(n) for one period (symbol section length Ts) of the spreading signal from instant mxe2x88x92N+1 to instant m so as to determine the correlation signal R(m) at instant m as the cross-correlation function of the received signal D(m) and the spreading signals P(n).
Note that, the output of the integrator 65 at the instant other than instant m is the value in calculation of the cross-correlation function in progress, and the length of an integration section does not reach the symbol section length Ts, and therefore is called a partial cross-correlation function. In the case where the spread ratio N is significantly large, the length of integration section may be made shorter than the symbol section length Ts, and the partial cross-correlation function is used as the correlation signal R(m).
The integrator 65 sets the integrated value which had been accumulated before the start of integration to zero. This operation of making the integrated value to zero is called damping or reset.
In order to carry out the inverse spread process with this correlator, unlike the above matched filter, it is required beforehand to make a coincidence of the phases of the received signal D(m) and the spreading signal P(n). Thus, the phase i of the spreading signal P(n) at the start of integration is controlled by the controller 67. The method using the correlator is called the active correlation method because the operation of actively matching the phases of the received signal D(m) and the spreading signal P(n) is required. Note that, instead of the integrator 65 of the correlator, a low-pass filter may be used.
The operation of actively matching the phases of the received signal and the spreading signal is the synchronism acquisition. In the matched filter, the synchronism acquisition can be carried out with the period equal to the symbol section length Ts. On the other hand, in the correlator, the controller 67 determines the integration result (correlation function) with respect to all the phase shift by shifting the phase i of the spreading signal P(n) at the start of integration by one per finish of one integration.
The synchronism acquisition time (time required for synchronism acquisition) of the correlator is the product of period N of the spreading signal P(n) and the integration time, and since the integration time is equal to the symbol section length Ts, the synchronism acquisition time is equal to Nxc3x97Ts. This is N times the synchronism acquisition time of the matched filter. As easily expected, by reducing the integration time shorter than the symbol section length Ts, the synchronism acquisition time can be reduced. However, this makes the output of the correlator the partial cross-correlation function, and as the integration time is reduced, the probability of synchronism acquisition failure is increased as a result of decrease in SN ratio of the signal outputted from the correlator. Thus, the integration time cannot be reduced without taking this problem into account.
Note that, such a synchronism acquisition using one correlator is called a serial search, and the correlator used here is called a sliding correlator.
As described, one characteristic of the matched filter is that the synchronism acquisition time is short. However, the matched filter has the problem that the circuit size tends to be large. A digital matched filter in particular poses this problem by the size of the adder 63, which is a multi-input adder. The multi-input adder can only be realized by the combination of two-input adders. Thus, when the tap number is N, at least Nxe2x88x921 two-input adders are required, and for this reason as the tap number is increased, the circuit size of the adder 63 is also increased. Also, because a high-speed operation is required as the chip section length Tc becomes shorter, the problem of increased power consumption is presented.
To solve these problems, an analog matched filter using an inverting amplifier has been catching attention. An example of this matched filter is disclosed in Japanese Unexamined Patent Publication No. 46231/1997 (Tokukaihei 9-46231) (published date Feb. 14, 1997). However, in the matched filter of this invention, when the tap number N becomes large, the problem of increased circuit size of the adder is presented.
FIG. 8 shows a schematic diagram of the adder disclosed in the above publication. In this adder, the number of inputs is N. Each input Vin(n) (n=0, 1, . . . , Nxe2x88x921) is connected to the input of the inverter amplifier AMP1 via a capacitor C(n). The output of the inverter amplifier AMP1 is feedbacked to the input of the inverter amplifier AMP1 via a capacitor Cf1. Thus, the output voltage Vout has the value which is the sum of inputs Vin(n) amplified by (c(n)/cf1). Note that, c(n) indicates the capacitance of the capacitor C(n) and cf1 indicates the capacitance of the capacitor Cf1.
Also, in the above adder, in order to have the function of the multiplier 62(n) of FIG. 6, two signals corresponding to the polarity of the spreading signals are added in advance, and the difference between these signals added together is determined. Also, because the number of inputs of the adder is limited by the driving power of an internal amplifier of the inverting amplifier, to constitute the adder 63 of FIG. 6, the adder of FIG. 8 needs to be used in plurality. Also, in the case that the analog matched filter is realized by LSI, it is required to make the capacitors of the adder large to some degree to reduce the adverse effects of various parasitic capacitance. Particularly, when the tap number is increased, the number N of the input capacitor C(n) of the amplifier AMP1 is also increased, and as a result the circuit size of the analog matched filter is increased. Further, when the tap number is large, the number Nxe2x88x921 of the delay elements 61(p) is also increased.
In each delay element 61(p), a sample/hold circuit is used. Each sample/hold circuit requires a capacitor for storing charge, and at least one amplifier for outputting the stored charge as a voltage without discharge. As an example, FIG. 9 schematically shows the sample/hold circuit used in the invention of the above publication.
The sample/hold circuit has an input as the inputs of the adder as shown in FIG. 8, and additionally a switch SW1. The input resistance of an inverting amplifier AMP 2 is extremely high so that the current flowing into the input of the amplifier AMP2 is negligible.
While the switch SW1 is closed (in sample section), input signal Vin is amplified by (xe2x88x92Ci2/Cf2) by an inverting amplifier composed of the amplifier AMP2 and the capacitors Ci2 and Cf2 and the amplified input signal Vin is outputted as output signal Vout from the inverting amplifier. While the switch SW1 is open (in hold section), the input signal Vin immediately before the switch SW1 opens is kept held by the capacitor Ci2, and thus (xe2x88x92Ci2/Cf2) times the input signal Vin is kept outputted as output signal Vout. When the analog matched filter is realized by LSI, as with the adder of FIG. 8, the capacitors Ci2 and Cf2 of the sample/hold circuit also need to have large capacitance. Therefore, the circuit size of the analog matched filter is also increased as a result.
Further, in the analog circuit, the settling time also presents a problem. As described, the received signal D(m) is sampled with respect to time. FIG. 11 shows an example of a waveform which has been sampled with respect to time. The ordinate and abscissa represents time and amplitude, respectively. The scale and numbers on ordinate indicate sampling instant. The sampling period is Tc. The waveform indicated by the solid line is the waveform with ideal sampling with respect to time, and a constant amplitude value is maintained between sampling instants. In the actual analog circuit, it takes some time to charge and discharge the capacitor. Thus, even when the input signal is ideal, the output cannot keep up with the abrupt amplitude change at the sampling instant, and it takes some time before the output reached a constant amplitude value, as shown by the waveform indicated by the broken line. The time it takes from the sampling instant to the time at which the output reaches the constant amplitude value is the settling time. The output of the adder 63 is the cross-correlation function so that it changes dynamically per instant. Namely, the problem of long settling time is presented by the fact that the amplitude fluctuation is large. When the settling time becomes longer than the sampling period length Tc, the matched filter cannot output accurate values.
To prevent this problem, the method of reducing the settling time of the adder of FIG. 8 and the sample/hold circuit of FIG. 9 comes to light. However, this method increases the current consumed by the amplifier. This is caused by the fact that the slew rate, which basically determines the settling time, is proportional to the bias current of the amplifier. The larger the slew rate, the shorter the settling time. The bias current of the amplifier becomes the current consumed by the amplifier. Thus, to reduce the settling time, it is required to increase the current consumed by the amplifier. The matched filter by design includes large numbers of delay elements and amplifiers in the adder constituting the matched filter, and thus when the current consumed by the amplifier is increased, the current consumed by the entire matched filter is increased notoriously.
Meanwhile, in the correlator, the degree of the adverse effect the magnitude of the spread ratio has on the circuit size is far smaller than the case of the matched filter. Yet, in a serial search which uses the correlator for synchronism acquisition, the problem of long acquisition time remains. To solve this problem, a parallel search which uses a plurality of correlators connected in parallel (parallel correlators) is available.
Japanese Unexamined Patent Publication No. 88526/1991 (Tokukaihei 3-88526) (published date Apr. 12, 1991) and Japanese Unexamined Patent Publication No. 125668/1989 (Tokukaihei 1-125668) (published date May 18, 1989) disclose this parallel search. In the technique disclosed in these publications, each of the parallel correlators carries out serial search, and for this reason the acquisition time is reduced by the inverse of the number of correlators provided, compared with the case of the serial search which uses only one correlator. However, as with the serial search, the parallel search is the active correlation method. Therefore, there is a problem that it is required to provide means for controlling the phase of the spreading signal inputted to the correlator.
Also, in the case of realizing the correlator by the analog circuit, the damping time may pose a problem.
In the output of the integrator of the correlator, the output fluctuation per instant is small, and accordingly the settling time is short. This is because the amplitude of the output of the integrator when the phases of the spreading signal and received signal coincide is simply increased or reduced as a partial cross-correlation function, and its absolute value takes the maximum value at the time the cross-correlation function is outputted. This maximum value is required to be a level which does not saturate as the integrator output, and therefore the increment (decrement) of the integrator output per instant need to be reduced.
On the other hand, in damping, the integration result, which became maximum when the phases of the spreading signal and received signal coincide, needs to be set to zero, and for this reason the settling time becomes long. In the serial search, in general, a single search is carried out in one symbol section Ts including this long damping time.
Thus, while the integration section can be reduced, it is prone to the problem that the SN ratio of the correlator output is reduced.
As disclosed in Japanese Unexamined Patent Publication No. 181345/1989 (Tokukaihei 1-181345) (published date Jul. 19, 1989), the method in which two correlators are used, and in which while one correlator carries out integration and the other correlator carries out damping has been available. However, in the parallel search using parallel correlators, when each correlator is realized by two correlators, the problem of large circuit size is presented. As described above, to reduce the settling time, the current consumed by the amplifier in the integrator is increased. However, because the number of amplifiers is large in parallel correlators, the problem is presented that the entire current consumed is increased notoriously.
An object of the present invention is to provide a matched filter with a small circuit size and low power consumption.
In order to achieve the above object, a matched filter of the present invention for inversely spreading a received signal using spreading signals in a spread spectrum communications receiver includes: a plurality of correlators for determining cross-correlation functions of the same received signal which has been sampled at a certain timing and spreading signals having a certain section length; a spreading signal delaying section for successively transferring the spreading signals having a certain section length with respect to the plurality of correlators by delaying a timing of transfer by a period equal to the section length of the spreading signals; and a selecting section for successively selecting one of the cross-correlation functions outputted from the plurality of correlators by a period equal to a sampling interval of the received signal.
In the matched filter of the present invention, the plurality of correlators operate in parallel, and to each of the plurality of correlators from the spreading signal delaying section are inputted spreading signals of different phases which are shifted from one another by a period equal to the section length of the spreading signal. Thus, the phases of the received signal and spreading signal coincide in one of the plurality of correlators.
The spreading signal delaying section for supplying the plurality of spreading signals of different phases can be realized by simple delay circuits, and it is not required to provide a circuit for controlling the phase of the spreading signals, which is necessary in the active correlation method such as a parallel search using parallel correlators, thus reducing circuit size.
In the case of realizing the matched filter of the present invention by an analog circuit, unlike the conventional matched filter of a transversal type, it is required to provide the selecting section, such as a multiplexer, which selects one input from plural inputs. However, because a multi-input adder is not required, the circuit size can be reduced by the size of this circuit.
Further, by increasing the number of correlators larger than a spread ratio, the damping time can be extended, thereby reducing power consumption.
Also, in the matched filter of the present invention, by extending the integration time, the cumulative process of the matched filter output can be carried out without a post-process which is required in a conventional matched filter.
Furthermore, as with the conventional matched filter of a transversal type, the matched filter of the present invention is applicable to an oversampled received signal, a long code spreading signal, and switching of the spreading signal, and therefore is capable of replacing the conventional matched filter of a transversal type in virtually all applications such a matched filter is used.
For a fuller understanding of the nature and advantages of the invention, reference should be made to the ensuing detailed description taken in conjunction with the accompanying drawings.